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19-2992; Rev 0; 10/03
ILABLE N KIT AVA EVALUATIO
-QSOP 16
3A, 1MHz, 1% Accurate, Internal Switch Step-Down Regulator with Power-OK
m
The MAX8505 step-down regulator operates from a 2.6V to 5.5V input and generates an adjustable output voltage from 0.8V to 0.85 VIN at up to 3A. With a 2.6V to 5.5V bias supply, the input voltage can be as low as 2.25V. The MAX8505 integrates power MOSFETs and operates at 1MHz/500kHz switching frequency to provide a compact design. Current-mode pulse-widthmodulated (PWM) control simplifies compensation with ceramic or polymer output capacitors and provides excellent transient response. The MAX8505 features 1% accurate output over load, line, and temperature variations. Adjustable soft-start is achieved with an external capacitor. During the soft-start period, the voltage-regulation loop is active. This limits the voltage dip when the active devices, such as microprocessors or ASICs connected to the MAX8505's output, apply a sudden load current step upon passing their undervoltage thresholds. The MAX8505 features current-limit, short-circuit, and thermal-overload protection and enables a rugged design. Open-drain power-OK (POK) monitors the output voltage.
P/ASIC/DSP/FPGA Core and I/O Supplies Chipset Supplies Server, RAID, and Storage Systems Network and Telecom Equipment
TOP VIEW
LX 1 IN 2 LX 3 IN 4 BST 5 VCC 6 POK 7 CTL 8 16 LX 15 PGND 14 LX
INPUT 2.6V TO 5.5V BST IN LX OUTPUT 0.8V TO 0.85 x VIN 3A
For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at 1-888-629-4642, or visit Maxim's website at www.maxim-ic.com.
6m
m x .9 m 4
General Description
Features
o Saves Space--4.9mm x 6mm Footprint, 1H Inductor, 47F Ceramic Output Capacitor o Input Voltage Range 2.6V to 5.5V Down to 2.25V with Bias Supply o 0.8V to 0.85 VIN, 3A Output o Ceramic or Polymer Capacitors o 1% Output Accuracy Over Load, Line, and Temperature o Fast Transient Response o Adjustable Soft-Start o In-Regulation Soft-Start Limits Output-Voltage Dips at Power-On o POK Monitors Output Voltage
MAX8505
Ordering Information
PART MAX8505EEE TEMP RANGE -40C to +85C PIN-PACKAGE 16 QSOP
Applications
Functional Diagram appears at end of data sheet.
Pin Configuration
Typical Operating Circuit
MAX8505
13 PGND 12 GND 11 REF 10 FB 9 COMP
ENABLE POWER-OK
MAX8505
VCC PGND FB COMP CTL REF POK GND
QSOP
________________________________________________________________ Maxim Integrated Products
1
3A, 1MHz, 1% Accurate, Internal Switch Step-Down Regulator with Power-OK MAX8505
ABSOLUTE MAXIMUM RATINGS
CTL, FB, IN, VCC to GND .........................................-0.3V to +6V COMP, REF, POK to GND ..........................-0.3V to (VCC + 0.3V) BST to LX..................................................................-0.3V to +6V PGND to GND .......................................................-0.3V to +0.3V Continuous Power Dissipation (TA = +70C) 16-Pin QSOP (derate 12.5mW/C above +70C).......1000mW Operating Temperature Range MAX8505EEE...................................................-40C to +85C Storage Temperature Range .............................-65C to +150C Junction Temperature ......................................................+150C Lead Temperature (soldering, 10s) .................................+300C
Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(VIN = VCC = VCTL = +3.3V, VFB = 0.8V, VCOMP = 1.25V, CREF = 0.01F, TA = 0C to +85C, unless otherwise noted.)
PARAMETER IN AND VCC IN Voltage Range VCC Voltage Range IN Supply Current VCC Supply Current Total Shutdown Current into IN and VCC VCC Undervoltage Lockout Threshold REF REF Voltage REF Shutdown Resistance REF Soft-Start Current Soft-Start Ramp Time FB FB Regulation Voltage FB Input Bias Current Maximum Output Current FB Threshold for POK Transition FB to POK Delay COMP COMP Transconductance Gain from FB to COMP From FB to COMP VCOMP = 1.25V to 1.75V 60 100 80 160 S dB IOUT_MAX VIN = 2.6V to 5.5V VFB = 0.7V VIN = VCC = 3.3V, VOUT = 1.2V, L = 1H/5.9m (Note 1) FB rising or falling FB rising or falling FB high FB low 3 10.5 -13.5 12 -12 50 13.5 -10.5 0.792 0.800 0.01 0.808 0.1 V A A % s VREF IREF = 0A, VIN = VCC = 2.6V to 5.5V From REF to GND, VCTL = 0V VREF = 0.4V Output from 0% to 100%, CREF = 0.01F to 1F 20 0.792 0.800 13 25 32 0.808 100 30 V A ms/F VIN VCC IIN ICC ISHDN UVLOth Switching with no load Switching with no load VIN = 3.3V VIN = 5.5V VCC = 3.3V VCC = 5.5V 2.25 2.6 6 10 3 6 20 2.40 2.2 2.35 50 2.55 10 VCC 5.5 10 V V mA mA A V SYMBOL CONDITIONS MIN TYP MAX UNITS
VIN = VCC = VBST - VLX = 5.5V, VCTL = 0V, VLX = 0 When LX starts/stops switching VCC rising VCC falling
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3A, 1MHz, 1% Accurate, Internal Switch Step-Down Regulator with Power-OK
ELECTRICAL CHARACTERISTICS (continued)
(VIN = VCC = VCTL = +3.3V, VFB = 0.8V, VCOMP = 1.25V, CREF = 0.01F, TA = 0C to +85C, unless otherwise noted.)
PARAMETER COMP Clamp Voltage, Low COMP Clamp Voltage, High COMP Shutdown Resistance LX (All LX Outputs Connected Together) LX On-Resistance, High LX On-Resistance, Low LX Current-Sense Transresistance LX Current-Limit Threshold LX Leakage Current LX Switching Frequency LX Minimum Off-Time LX Maximum Duty Cycle LX Minimum Duty Cycle SLOPE COMPENSATION Slope Compensation BST BST Shutdown Supply Current CTL For 1MHz CTL Input Threshold CTL Input Current POK (Power-OK) POK Output Voltage, Low POK Leakage Current POK Fault Delay Time THERMAL SHUTDOWN Thermal-Shutdown Threshold Thermal-Shutdown Hysteresis When LX stops switching TJ rising +170 20 C C VFB = 0.6V or 1.0V, IPOK = 2mA VPOK = 5.5V From FB to POK, any threshold 25 25 0.001 50 100 1 100 mV A s VIN = VCC = 2.6V, 3.3V, 5.5V For 500kHz For shutdown VCTL = 0V or 5.5V, VIN = VCC = 5.5V -1 80 55 70 45 +1 % of VCC A (VBST - VLX) = VIN = VCC = 5.5V, VCTL = 0 VLX = 5.5V VLX = 0V LX open 10 10 10 A Extrapolated to 100% duty cycle 245 300 400 mV RT VIN = VBST - VLX = 3.3V VIN = VBST - VLX = 2.6V VIN = VBST - VLX = 3.3V VIN = VBST - VLX = 2.6V From LX to COMP Sourcing, Typical Application Circuit Sinking, VIN = VCC = 2.6V to 5.5V VIN = VCC = 5.5V, VCTL = 0 VIN = VCC = 2.6V, 3.3V, 5.5V LX = 5.5V LX = 0V CTL = VCC CTL = 2/3VCC 500kHz 1MHz 500kHz 1MHz -100 0.85 0.44 95 90 84 1 0.5 110 94 89 5 10 8 15 1.15 0.56 135 0.068 4.6 -4.3 38 42 38 42 0.086 5.6 -2.6 0.104 6.6 -1.0 100 74 74 m m A A MHz ns % % SYMBOL CONDITIONS VIN = VCC = 2.6V, 3.3V, 5.5V, VFB = 0.9V VIN = VCC = 2.6V, 3.3V, 5.5V, VFB = 0.7V From COMP to GND, VCTL = 0V MIN 0.45 1.7 TYP 0.75 1.9 13 MAX 1.00 2.1 100 UNITS V V
MAX8505
VIN = VCC = 2.6V, 3.3V, 5.5V VIN = VCC = 2.6V, 3.3V, 5.5V VIN = VCC = 2.6V, 3.3V, 5.5V
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3A, 1MHz, 1% Accurate, Internal Switch Step-Down Regulator with Power-OK MAX8505
ELECTRICAL CHARACTERISTICS
(VIN = VCC = VCTL = +3.3V, VFB = 0.8V, VCOMP = 1.25V, CREF = 0.01F, TA = -40C to +85C, unless otherwise noted.) (Note 2)
PARAMETER IN AND VCC IN Voltage Range VCC Voltage Range IN Supply Current VCC Supply Current Total Shutdown Current into IN and VCC VCC Undervoltage Lockout Threshold REF REF Voltage REF Shutdown Resistance REF Soft-Start Current FB FB Regulation Voltage FB Input Bias Current Maximum Output Current FB Threshold for POK Transition COMP COMP Transconductance COMP Clamp Voltage, Low COMP Clamp Voltage, High COMP Shutdown Resistance LX (All LX Outputs Connected Together) LX On-Resistance, High LX On-Resistance, Low LX Current-Sense Transresistance LX Current-Limit Threshold LX Leakage Current LX Switching Frequency LX Minimum Off-Time RT VIN = VBST - VLX = 3.3V VIN = VBST - VLX = 3.3V From LX to COMP Sourcing, Typical Application Circuit Sinking, VIN = VCC = 2.6V to 5.5V VIN = VCC = 5.5V, VCTL = 0 VIN = VCC = 2.6V, 3.3V, 5.5V LX = 5.5V LX = 0V CTL = VCC CTL = 2/3 VCC -100 0.85 0.44 95 1.15 0.56 135 0.068 4.6 -4.3 74 74 0.104 5.6 -1.0 100 m m A A MHz ns From FB to COMP VIN = VCC = 2.6V, 3.3V, 5.5V, VFB = 0.9V VIN = VCC = 2.6V, 3.3V, 5.5V, VFB = 0.7V From COMP to GND, VCTL = 0V 60 0.45 1.7 160 1.00 2.1 100 S V V IOUT_MAX VFB VIN = 2.6V to 5.5V VFB = 0.7V VIN = VCC = 3.3V, VOUT = 1.2V, L = 1H/5.9m (Note 1) FB rising or falling FB high FB low 3 10.5 -13.5 13.5 -10.5 0.791 0.808 0.1 V A A % IIN ICC ISHDN UVLOth Switching with no load Switching with no load VIN = 3.3V VCC = 3.3V VIN 2.25 2.6 VCC 5.5 10 10 50 2.55 2.2 0.791 20 0.808 100 30 V V mA mA A V SYMBOL CONDITIONS MIN TYP MAX UNITS
VIN = VCC = VBST - VLX = 5.5V, VCTL = 0V, VLX = 0 When LX starts/stops switching VCC rising VCC falling
VREF
IREF = 0A, VIN = VCC = 2.6V to 5.5V From REF to GND, VCTL = 0V VREF = 0.4V
V A
VIN = VCC = 2.6V, 3.3V, 5.5V
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3A, 1MHz, 1% Accurate, Internal Switch Step-Down Regulator with Power-OK
ELECTRICAL CHARACTERISTICS (continued)
(VIN = VCC = VCTL = +3.3V, VFB = 0.8V, VCOMP = 1.25V, CREF = 0.01F, TA = -40C to +85C, unless otherwise noted.) (Note 2)
PARAMETER LX Maximum Duty Cycle LX Minimum Duty Cycle SLOPE COMPENSATION Slope Compensation BST VLX = 5.5V BST Shutdown Supply Current CTL For 1MHz CTL Input Threshold CTL Input Current POK (Power-OK) POK Output Voltage, Low POK Leakage Current POK Fault Delay Time VFB = 0.6V or 1.0V, IPOK = 2mA VPOK = 5.5V From FB to POK, any threshold 25 100 1 100 mV A s VIN = VCC = 2.6V, 3.3V, 5.5V For 500kHz For shutdown VCTL = 0V or 5.5V, VIN = VCC = 5.5V -1 80 55 70 45 +1 % of VCC A (VBST - VLX) = VIN = VCC = 5.5V, VCTL = 0 VLX = 0V LX open 10 10 10 A Extrapolated to 100% duty cycle 245 406 mV SYMBOL CONDITIONS VIN = VCC = 2.6V, 3.3V, 5.5V VIN = VCC = 2.6V, 3.3V, 5.5V 500kHz 1MHz 500kHz 1MHz MIN 90 84 8 15 TYP MAX UNITS % %
MAX8505
Note 1: Under normal operating conditions, COMP moves between 1.25V and 2.15V as the duty cycle changes from 10% to 90% and peak inductor current changes from 0 to 3A. Maximum output current is related to peak inductor current, inductor value input voltage, and output voltage by the following equations:
IOUT _ MAX = ILIM - (1 - D) x t S x VOUT / 2L 1 + (1 - D) x t S x (RNLS + RL ) / 2L
where VOUT = output voltage; ILIM = current limit of high-side switch; tS = switching period; RL = ESR of inductor; RNLS = on-resistance of low-side switch; L = inductor. Equations for ILIM and D are shown as follows:
ILIM = ILIM _ DC100 + VSW 1- D RT
where ILIM_DC100 = current limit at D = 100%; RT = transresistance from LX to COMP; VSW = slope compensation (310mV 20%); D = duty cycle:
V + I (R + RL ) D = OUT O NLS VIN + IO (RNLS - RNHS )
where VOUT = output voltage; VIN = input voltage; IO = output current; RL = ESR of inductor; RNHS = on-resistance of highside switch; RNLS = on-resistance of low-side switch. See the Typical Application Circuit for external components. Note 2: Specifications to -40C are guaranteed by design and not production tested. Note 3: LX has internal clamp diodes to PGND and IN pins 2 and 4. Applications that forward bias these diodes should take care not to exceed the IC's package power dissipation limits. Note 4: When connected together, the LX output is designed to provide 3.5ARMS current.
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3A, 1MHz, 1% Accurate, Internal Switch Step-Down Regulator with Power-OK MAX8505
Typical Operating Characteristics
(Typical values are at VIN = VCC = VCTL = 5V, VOUT = 1.2V, IOUT = 3A, and TA = +25C, unless otherwise noted.)
EFFICIENCY vs. OUTPUT CURRENT (VIN = VCC = 5V, fSW = 1MHz)
MAX8505 toc01
EFFICIENCY vs. OUTPUT CURRENT (VIN = VCC = 3.3V, fSW = 500kHz)
MAX8505 toc02
EFFICIENCY vs. OUTPUT CURRENT (VIN = VCC = 3.3V, fSW = 1MHz)
A 90 B EFFICIENCY (%)
MAX8505 toc03
100 A 90 B EFFICIENCY (%) C 80 D
100
A B D C
100
90 EFFICIENCY (%)
80
80 D 70 A: VOUT = 2.5V B: VOUT = 1.8V C: VOUT = 1.2V D: VOUT = 0.8V 0 1 2 OUTPUT CURRENT (A) 3
C
70 A: VOUT = 3.3V B: VOUT = 2.5V C: VOUT = 1.2V D: VOUT = 0.8V 0 1 2 OUTPUT CURRENT (A) 3 4
70 A: VOUT = 2.5V B: VOUT = 1.8V C: VOUT = 1.2V D: VOUT = 0.8V 0 1 2 OUTPUT CURRENT (A) 3 4
60
60
60
50
50
50 4
EFFICIENCY vs. OUTPUT CURRENT (VIN = 2.5V, VCC = 5V, fSW = 1MHz)
MAX8505 toc04
FREQUENCY vs. INPUT VOLTAGE AND TEMPERATURE
MAX8505 toc05
100 A 90 B EFFICIENCY (%) 80 C
1.05
1.03 FREQUENCY (MHz) +85C 1.01 +25C
70
0.99
60
A: VOUT = 1.8V B: VOUT = 1.2V C: VOUT = 0.8V 0 1 2 OUTPUT CURRENT (A) 3 4
0.97
-40C
50
0.95 2.5 3.0 3.5 4.0 4.5 5.0 5.5 INPUT VOLTAGE (V)
FREQUENCY vs. INPUT VOLTAGE AND TEMPERATURE
MAX8505 toc06
OUTPUT LOAD REGULATION
A: VOUT = 0.8V B: VOUT = 1.2V C: VOUT = 1.8V D: VOUT = 2.5V D 3 2 1 VIN = VCC = 3.3V 0 C B A
MAX8505 toc07
530 520 FREQUENCY (kHz) 510 +25C 500 490 480 470 2.5 3.0 3.5 4.0 4.5 5.0 +85C
6 5 4
-40C
5.5
-VOUT (mV)
0
1
2 OUTPUT CURRENT (A)
3
4
INPUT VOLTAGE (V)
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3A, 1MHz, 1% Accurate, Internal Switch Step-Down Regulator with Power-OK
Typical Operating Characteristics (continued)
(Typical values are at VIN = VCC = VCTL = 5V, VOUT = 1.2V, IOUT = 3A, and TA = +25C, unless otherwise noted.)
MAX8505
SHUTDOWN SUPPLY CURRENT vs. INPUT VOLTAGE
MAX8505 toc08
CURRENT LIMIT vs. OUTPUT VOLTAGE
5.4 5.3 CURRENT LIMIT (A) 5.2 5.1 5.0 4.9 4.8 4.7 fSW = 1MHz
MAX8505 toc09
0.4 SHUTDOWN SUPPLY CURRENT (mA)
5.5
0.3
0.2
0.1
0 2.5 3.0 3.5 4.0 4.5 INPUT VOLTAGE (V)
fSW = 1MHz 5.0 5.5
4.6 4.5 0.8 1.3 1.8 2.3 2.8 3.3 OUTPUT VOLTAGE (V)
OUTPUT SHORT-CIRCUIT CURRENT vs. INPUT VOLTAGE
fSW = 1MHz
MAX8505 toc10
GND-MEASURED TEMPERATURE vs. OUTPUT CURRENT
GND-MEASURED TEMPERATURE (C) TA = +85C 100 TA = +25C 80 60 40 20 TA = -40C 0
MAX8505 toc11
5.5 OUTPUT SHORT-CIRCUIT CURRENT (A)
120
4.5
3.5
VIN = 5V, VOUT = 1.5V
2.5 2.5 3.0 3.5 4.0 4.5 5.0 5.5 INPUT VOLTAGE (V)
3.00
3.25
3.50
3.75
4.00
OUTPUT CURRENT (A)
REFERENCE VOLTAGE vs. TEMPERATURE
MAX8505 toc12
TRANSIENT RESPONSE (VIN = 5V, VOUT = 1.2V)
MAX8505 toc13
0.810
REFERENCE VOLTAGE (V)
0.805
OUTPUT VOLTAGE AC-COUPLED 100mV/div
0.800 2.25A 0.795 fSW = 1MHz 0.790 -40 -15 10 35 60 85 110 40s/div TEMPERATURE (C) OUTPUT 0.75A CURRENT 1A/div 0
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3A, 1MHz, 1% Accurate, Internal Switch Step-Down Regulator with Power-OK MAX8505
Typical Operating Characteristics (continued)
(Typical values are at VIN = VCC = VCTL = 5V, VOUT = 1.2V, IOUT = 3A, and TA = +25C, unless otherwise noted.)
TRANSIENT RESPONSE (VIN = 3.3V, VOUT = 1.2V)
MAX8505 toc14
SWITCHING WAVEFORM (VIN = 5V, VOUT = 1.2V, IOUT = 2.5A)
MAX8505 toc15
OUTPUT VOLTAGE AC-COUPLED 100mV/div
VLX 2V/div
2.25A OUTPUT 0.75A CURRENT 1A/div 0
INDUCTOR CURRENT AC-COUPLED 2A/div VOUT AC-COUPLED 20mV/div 200ns/div
40s/div
SOFT-START/SHUTDOWN WAVEFORM (VIN = 3.3V, VOUT = 1.2V, IOUT = 3A, CREF = 0.068F)
MAX8505 toc16
TRANSIENT RESPONSE DURING SOFT-START
MAX8505 toc17
VOUT 500mV/div VCTRL 5V/div INPUT CURRENT 1A/div
VOUT 100mV/div
IOUT 2A/div VPOK 5V/div 400s/div 100s/div
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3A, 1MHz, 1% Accurate, Internal Switch Step-Down Regulator with Power-OK
Pin Description
PIN 1, 3, 14, 16 2, 4 5 6 7 8 9 10 11 12 13, 15 NAME LX IN BST VCC POK CTL COMP FB REF GND PGND FUNCTION Inductor Connection. Connect an inductor between these pins and the regulator output. All LX pins must be connected together externally. Connect a 3300pF ceramic capacitor from LX to PGND. Power-Supply Inputs. Ranges from 2.6V to 5.5V. Bypass with two ceramic 22F capacitors to GND. All IN pins must be connected together externally. Bootstrapped Voltage Input. High-side driver supply pin. Bypass to LX with a 0.1F capacitor. Charged from IN with an external Schottky diode. Supply Voltage and Gate-Drive Supply for Low-Side Driver. Decouple with a 10 resistor and bypass to GND with 0.1F. Power-OK Output. Open-drain output of a window comparator that pulls POK low when the FB pin is outside the 0.8V 12% range. Output Control. When at GND, the regulator is off. When at VCC, the regulator is operating at 1MHz. For a 500kHz application, raise the pin to 2/3 VCC. Regulator Loop Compensation. Connect a series RC network to GND. This pin is pulled to GND when the output is shut down, or in UVLO or thermal shutdown. Feedback Input. This pin regulates to 0.8V. Use an external resistive-divider from the output to set the output voltage. Place a capacitor at this pin to set the soft-start time. This pin goes to 0V when the part is shut down. Ground Power Ground. Connect this pin to GND at a single point.
MAX8505
Detailed Description
The MAX8505 is a high-efficiency synchronous buck regulator capable of delivering up to 3A of output current. It operates in PWM mode at a high fixed frequency of 500kHz or 1MHz, thereby reducing external component size. The MAX8505 operates from a 2.6V to 5.5V input voltage and can produce an output voltage from 0.8V to 0.85 VIN.
Controller Block Function
The MAX8505 step-down converter uses a PWM current-mode control scheme. An open-loop comparator compares the voltage-feedback error signal against the sum of the amplified current-sense signal and the slope compensation ramp. At each rising edge of the internal clock, the internal high-side MOSFET turns on until the PWM comparator trips. During this on-time, current ramps up through the inductor, sourcing current to the output and storing energy in the inductor. The current-mode feedback system regulates the peak inductor current as a function of the output-voltage error signal. Since the average inductor current is nearly the same as the peak inductor current, the circuit acts as a switch-mode transconductance amplifier. To preserve inner-loop stability and eliminate inductor staircasing, a slope-
compensation ramp is summed into the main PWM comparator. During the second half of the cycle, the internal high-side N-channel MOSFET turns off, and the internal low-side N-channel MOSFET turns on. The inductor releases the stored energy as its current ramps down while still providing current to the output. The output capacitor stores charge when the inductor current exceeds the load current, and discharges when the inductor current is lower, smoothing the voltage across the load. Under overload conditions, when the inductor current exceeds the current limit (see the Current Limit section), the high-side MOSFET does not turn on at the rising edge of the clock and the low-side MOSFET remains on to let the inductor current ramp down.
Current Sense
An internal current-sense amplifier produces a current signal proportional to the voltage generated by the highside MOSFET on-resistance and the inductor current (RDS(ON) ILX). The amplified current-sense signal and the internal slope-compensation signal are summed together into the comparator's inverting input. The PWM comparator turns off the internal high-side MOSFET when this sum exceeds the feedback voltage from the voltage-error amplifier.
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3A, 1MHz, 1% Accurate, Internal Switch Step-Down Regulator with Power-OK MAX8505
Current Limit
The MAX8505 offers both high-side and low-side current limits. The high-side current limit monitors the inductor peak current and the low-side current limit monitors the inductor valley current. Current-limit thresholds are 6A (typ) for high side and 3.8A (typ) for low side. If the output inductor current exceeds the highside current limit during its on-time, the high-side MOSFET turns off and the synchronous rectifier turns on. The inductor current is continuously monitored during the on-time of the low-side MOSFET. If the inductor current is still above the low-side current limit at the moment of the next clock cycle, the high-side MOSFET is not turned on and the low-side MOSFET is kept on to continue discharging the output inductor current. Once the inductor current is below the low-side current limit, the high-side MOSFET is turned on at the next clock cycle. If the inductor current stays less than the high-side current limit during the minimum on duty ratio, the normal operation resumes at the next clock cycle. Otherwise, the current-limit operation continues.
Soft-Start
To reduce input transient currents during startup, a programmable soft-start is provided. The soft-start time is given by: t SOFT _ START = CREF x 0.8V 25A
A minimum capacitance of 0.01F at REF is recommended to reduce the susceptibility to switching noise.
Power-OK (POK)
The MAX8505 also includes an open-drain POK output that indicates when the regulator output is within 12% of its nominal output. If the output voltage moves outside this range, the POK output is pulled to ground. Since this comparator has no hysteresis on either threshold, a 50s delay time is added to prevent the POK output from chattering between states. The POK should be pulled to VIN or another supply voltage less than 5.5V through a resistor.
VCC Decoupling Due to the high switching frequency and tight output tolerance (1%), decouple V CC from IN with a 10 resistor and bypass to GND with a 0.1F capacitor. Place the capacitor as close to VCC as possible. Bootstrap (BST)
Gate-drive voltage for the high-side N-channel switch is generated by a bootstrapped capacitor boost circuit. The bootstrapped capacitor is connected between the BST pin and LX. When the low-side N-channel MOSFET is on, it forces LX to ground and charges the capacitor to VIN through diode D1. When the low-side N-channel MOSFET turns off and the high-side N-channel MOSFET turns on, LX is pulled to VIN. D1 prevents the capacitor from discharging into VIN and the voltage on the bootstrapped capacitor is boosted above VIN. This provides the necessary voltage for the high driver. A Schottky diode should be used for D1.
UVLO
If VCC drops below +2.25V, the UVLO circuit inhibits switching. Once VCC rises above +2.35V, the UVLO clears, and the soft-start sequence activates.
Thermal Protection
Thermal-overload protection limits total power dissipation in the device. When the junction temperature exceeds T J = +170C, a thermal sensor forces the device into shutdown, allowing the die to cool. The thermal sensor turns the device on again after the junction temperature cools by 20C, resulting in a pulsed output during continuous overload conditions. Following a thermal-shutdown condition, the soft-start sequence begins anew.
Design Procedure
Duty Cycle
The equation below shows how to calculate the resulting duty cycle when series losses from the inductor and internal switches are accounted for:
D= VOUT + IOUT (RNLS + RL ) VOUT + IOUT (RNLS + RL ) = VIN + IOUT (RNLS - RNHS ) VIN if RNLS = RNHS
Frequency Selection/Enable (CTL)
The MAX8505 includes a frequency selection circuit to allow it to run at 500kHz or 1MHz. The operating frequency is selected through a control input, CTL, which has three input threshold ranges that are ratiometric to the input supply voltage. When CTL is driven to GND, it acts like an enable pin, switching the output off. When the CTL input is driven to >0.8 VCC, the MAX8505 is enabled with 1MHz switching. When the CTL input is between 0.55 VCC and 0.7 VCC, the part operates at 500kHz. When the CTL input is <0.45 x VCC, the device is in shutdown.
10
where V OUT = output voltage; V IN = input voltage; IOUT = output current (3A maximum); RL = ESR of the inductor; RNHS = on-resistance of the high-side switch; and RNLS = on-resistance of the low-side switch.
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3A, 1MHz, 1% Accurate, Internal Switch Step-Down Regulator with Power-OK
Output Voltage Selection
The output voltage of the MAX8505 can be adjusted from 0.8V to 85% of the input voltage at 500kHz or up to 80% of the input voltage at 1MHz. This is done by connecting a resistive-divider (R2 and R3) between the output and the FB pin (see the Typical Operating Circuit). For best results, keep R3 below 50k and select R2 using the following equation: V R2 = R3 x OUT - 1 VREF where VREF = 0.8V.
VRIPPLE = VRIPPLE(C)2 + VRIPPLE(ESR)2 + VRIPPLE(ESL)2
MAX8505
where the output ripples due to output capacitance, ESR, and ESL are: VRIPPLE(C) = IP - P 8 x COUT x fS
VRIPPLE(ESR) = IP-P x ESR I I VRIPPLE(ESL) = P - P x ESL or P - P x ESL, t ON t OFF or, whichever is greater. The ESR is the main contribution to the output voltage ripple. IP-P, the peak-to-peak inductor current, is: IP - P = (VIN - VOUT ) VOUT x fS x L VIN
Inductor Design
When choosing the inductor, the key parameters are inductor value (L) and peak current (I PEAK ). The following equation includes a constant, denoted as LIR, which is the ratio of peak-to-peak inductor AC current (ripple current) to maximum DC load current. A higher value of LIR allows smaller inductance but results in higher losses and ripple. A good compromise between size and losses is found at approximately 20% to 30% ripple-current to load-current ratio (LIR = 0.20 to 0.30): V x (1 - D) L = OUT IOUT x LIR x fS where fS is the switching frequency and (I -I ) LIR = 2 x PEAK OUT IOUT Choose an inductor with a saturation current at least as high as the peak inductor current. Additionally, verify the peak inductor current does not exceed the current limit. The inductor selected should exhibit low losses at the chosen operating frequency. Output Capacitor Design and Output Ripple The key selection parameters for the output capacitor are capacitance, ESR, ESL, and the voltage rating requirements. These affect the overall stability, output ripple voltage, and transient response of the DC-DC converter. The output ripple occurs due to variations in the charge stored in the output capacitor, the voltage drop due to the capacitor's ESR, and the voltage drop due to the capacitor's ESL. Calculate the output voltage ripple due to the output capacitance, ESR, and ESL as:
Use these equations for initial capacitor selection, but determine final values by testing a prototype or evaluation circuit. As a rule, a smaller ripple current results in less output voltage ripple. Since the inductor ripple current is a factor of the inductor value, the output voltage ripple decreases with larger inductance. Use ceramic capacitors for their low ESR and ESL at the switching frequency of the converter. The low ESL of ceramic capacitors makes ripple voltages negligible. Load-transient response depends on the selected output capacitor. During a load transient, the output instantly changes by ESR ILOAD. Before the controller can respond, the output deviates further, depending on the inductor and output capacitor values. After a short time (see Transient Response in the Typical Operating Characteristics), the controller responds by regulating the output voltage back to its nominal state. The controller response time depends on the closed-loop bandwidth, the inductor value, and the slew rate of the transconductance amplifier. A higher bandwidth yields a faster response time, thus preventing the output from deviating further from its regulating value.
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11
3A, 1MHz, 1% Accurate, Internal Switch Step-Down Regulator with Power-OK
Input Capacitor Design The input capacitor reduces the current peaks drawn from the input power supply and reduces switching noise in the IC. The impedance of the input capacitor at the switching frequency should be less than that of the input source so high-frequency switching currents do not pass through the input source but instead are shunted through the input capacitor. A high source impedance requires larger input capacitance. The input capacitor must meet the ripple current requirement imposed by the switching currents. The RMS input ripple current is given by: IRIPPLE = ILOAD x VOUT x (VIN - VOUT ) VIN2
MAX8505
For customized compensation networks that increase stability or transient response, the simplified loop gain can be described by the equation: AVOL = VFB x gmERR x ROERR x VOUT
s x CCOMP x RCOMP + 1 x (s x C x RCOMP + 1) x (s x CPARA x RCOMP + 1) COMP RL s x COUT x RESR + 1 x R T s x COUT x RL + 1 where: gmERR (COMP transconductance) = 100mho ROERR (output resistance of transconductance amplifier) = 20M CCOMP (compensation capacitor at COMP pin) RT (current-sense transresistance) = 0.086 CPARA (parasitic capacitance at COMP pin) = 10pF RL (load resistor) COUT (output capacitor) RESR (series resistance of COUT) s = j2f In designing the compensation circuit, select an appropriate converter bandwidth (fC) to stabilize the system while maximizing transient response. This bandwidth should not exceed 1/10 of the switching frequency. Use 100kHz as a reasonable starting point. Calculate CCOMP based on this bandwidth using the following equation: I x RT x (R3 + R2) x 2 x fC x COUT RCOMP = OUT VOUT x gmERR x R2 where R2 and R3 are the feedback resistors. Calculate CCOMP to cancel out the pole created by RL and COUT using the following equation; CCOMP = RL x COUT RCOMP
where IRIPPLE is the input RMS ripple current. Use sufficient input bypass capacitance to ensure that the absolute maximum voltage rating of the MAX8505 is not exceeded in any condition. When input supply is not located close to the MAX8505, a bulk bypass input capacitor may be needed. Compensation Design The double pole formed by the inductor and output capacitor of most voltage-mode controllers introduces a large phase shift, which requires an elaborate compensation network to stabilize the control loop. The MAX8505 controller utilizes a current-mode control scheme that regulates the output voltage by forcing the required current through the external inductor, eliminating the double pole caused by the inductor and output capacitor, and greatly simplifying the compensation network. A simple type 1 compensation with single compensation resistor (R1) and compensation capacitor (C8) create a stable and high-bandwidth loop (see the Typical Operating Circuit). An internal transconductance error amplifier compensates the control loop. Connect a series resistor and capacitor between COMP (the output of the error amplifier) and GND to form a pole-zero pair. The external inductor, internal current-sensing circuitry, output capacitor, and external compensation circuit determine the loop stability. Choose the inductor and output capacitor based on performance, size, and cost. Additionally, select the compensation resistor and capacitor to optimize controlloop stability. The component values shown in the Typical Operating Circuit yield stable operation over a broad range of input-to-output voltages.
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3A, 1MHz, 1% Accurate, Internal Switch Step-Down Regulator with Power-OK
Applications Information
PC Board Layout Considerations
Careful PC board layout is critical to achieve clean and stable operation. The switching power stage requires particular attention. Follow these guidelines for good PC board layout: 1) Place decoupling capacitors as close to the IC as possible. Keep power ground plane (connected to PGND) and signal ground plane (connected to GND) separate. 2) Connect input and output capacitors to the power ground plane; connect all other capacitors to the signal ground plane. 3) Keep the high-current paths as short and wide as possible. Keep the path of switching current short and minimize the loop area formed by the high-side MOSFET, the low-side MOSFET, and the input capacitors. Avoid vias in the switching paths. 4) If possible, connect IN, LX, and PGND separately to a large copper area to help cool the IC to further improve efficiency and long-term reliability. 5) Ensure all feedback connections are short and direct. Place the feedback resistors as close to the IC as possible. 6) Route high-speed switching nodes away from sensitive analog areas (FB, COMP).
MAX8505
Chip Information
TRANSISTOR COUNT: 3352 PROCESS: BiCMOS
Typical Application Circuit
D1 (CENTRAL CMOSH-3) C7 0.1F L1 1H (FDV3H-IRON) LX C9 COUT 3300pF 47F (6.3V CERAMIC)
VIN 2.6V TO 5.5V
BST IN R7 10 C5 0.1F R6 20k VCC
VOUT 1.2V R2 11.3k
C2 2 x 22F (10V CERAMIC)
MAX8505
PGND FB COMP REF
R1 51k R3 22.6k
CTL POK
POWER-OK
GND
C6 0.01F
C8 220pF
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13
3A, 1MHz, 1% Accurate, Internal Switch Step-Down Regulator with Power-OK MAX8505
Functional Diagram
REFERENCE 1.25V
POK
MAX8505
50s N VCC
FB GND
UVLO
BST
IN 25A REF PWM 25A LX
FB
PGND
GND
COMP
CTL
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3A, 1MHz, 1% Accurate, Internal Switch Step-Down Regulator with Power-OK
Package Information
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information, go to www.maxim-ic.com/packages.)
QSOP.EPS
MAX8505
PACKAGE OUTLINE, QSOP .150", .025" LEAD PITCH
21-0055
E
1 1
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 ____________________ 15 (c) 2003 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products.

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